## Electronic dispersion compensation using full optical-field reconstruction in 10Gbit/s OOK based systems

Optics Express, Vol. 16, Issue 20, pp. 15353-15365 (2008)

http://dx.doi.org/10.1364/OE.16.015353

Acrobat PDF (250 KB)

### Abstract

We investigate the design of electronic dispersion compensation (EDC) using full optical-field reconstruction in 10Gbit/s on-off keyed transmission systems limited by optical signal-to-noise ratio (OSNR). By effectively suppressing the impairment due to low-frequency component amplification in phase reconstruction, properly designing the transmission system configuration to combat fiber nonlinearity, and successfully reducing the vulnerability to thermal noise, a 4.8dB OSNR margin can be achieved for 2160km single-mode fiber transmission without any optical dispersion compensation. We also investigate the performance sensitivity of the scheme to various system parameters, and propose a novel method to greatly enhance the tolerance to differential phase misalignment of the asymmetric Mach-Zehnder interferometer. This numerical study provides important design guidelines which will enable full optical-field EDC to become a cost-effective dispersion compensation solution for future transparent optical networks.

© 2008 Optical Society of America

## 1. Introduction

1. S. Schube and M. Mazzini, “Testing and interoperability of 10GBASE-LRM optical interfaces,” IEEE Commun. Mag. **45**, s26–s31 (2007). [CrossRef]

4. D. McGhan, M. O’Sullivan, M. Sotoodeh, A. Savchenko, C. Bontu, M. Belanger, and K. Roberts, “Electronic dispersion compensation,” in *Proc. Optical Fiber Communication Conference* (2006), paper OWK1. [CrossRef]

5. G. Bosco and P. Poggiolini, “Long-distance effectiveness of MLSE IMDD receivers,” IEEE Photo. Technol. Lett. **18**, 1037–1039 (2006). [CrossRef]

8. X. Liu, S. Chandrasekhar, and A. Leven, “Digital self-coherent detection”, Opt. Express **16**, 792–803 (2008). [CrossRef] [PubMed]

8. X. Liu, S. Chandrasekhar, and A. Leven, “Digital self-coherent detection”, Opt. Express **16**, 792–803 (2008). [CrossRef] [PubMed]

## 2. Principle and simulation model

*t*and differential phase shift of

*π*/2. Assuming that the input optical field (baseband representation) is

*E*(

*t*), the two outputs of the AMZI,

*E*

_{1}(

*t*) and

*E*

_{2}(

*t*), are given by:

*ω*(

*t*) (=

*ω*(

*t*)-

*ω*

_{0}) is the instantaneous frequency of baseband representation of the optical field,

*E*(

*t*).

*ω*

_{0}is the optical carrier frequency.

*E*

_{1}(

*t*) and

*E*

_{2}(

*t*) are detected by a pair of detectors, electrically amplified, and filtered by electrical filters (EFs) to obtain the electrical signals

*V*

_{1}(

*t*) and

*V*

_{2}(

*t*):

*α*

_{1}and

*α*

_{2}are scaling factors taking account of the responsivities of the detectors and the gains of the electrical amplifiers for the

*V*

_{1}(

*t*) and

*V*

_{2}(

*t*) signal paths. For analytical simplicity,

*α*

_{1}=

*α*

_{2}=1 is assumed. Signals proportional to the intensity, instantaneous frequency, and phase information of the optical field,

*V*

_{A}(

*t*),

*V*

_{f}(

*t*), and

*V*

_{p}(

*t*), can be extracted by signal processing of

*V*

_{1}(

*t*) and

*V*

_{2}(

*t*):

*ω*(

*t*)Δ

*t*)/(2πΔ

*t*)≈Δ

*ω*(

*t*)/2π is satisfied given Δ

*ω*(

*t*)Δ

*t*<<1. The intensity and frequency information is exploited to reconstruct a replica of the optical signal in the radio frequency (RF) range by applying the signals

*V*

_{A}and

*V*

_{f}to amplitude and frequency modulators respectively, allowing for subsequent full optical-field compensation using a dispersive transmission line [7]. This scheme features better compensation performance compared to conventional DD EDC as it benefits from the knowledge of the recovered phase information, and also better cost effectiveness compared to the coherent-detection based EDC by avoiding complicated phase and polarization tracking or recovery.

*V*

_{1}-

*V*

_{2}and an additional single photodiode to obtain

*V*

_{1}+

*V*

_{2}[11].

^{11}-1 pseudo-random binary sequence (PRBS) repeated nine times (18,423 bits). 10 ‘0’ bits and 11 ‘0’ bits were added before and after this data train respectively to simplify the boundary conditions. The bits were raised-cosine shaped with a roll-off coefficient of 0.4 and 40 samples per bit. The extinction ratio (ER) of the modulated OOK signal was set by adjusting the bias and the amplitude of the electrical OOK data.

*n*

_{sp}

*hν*(

*G*-1) for each polarization, where

*G*and

*hν*are the amplifier gain and the photon energy respectively.

*n*

_{sp}is population inversion factor of the amplifiers and was set to give 4dB amplifier noise figure (NF).

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*), as shown in Fig. 2. The AMZI for the extraction of

*V*

_{y}(

*t*) had

*π*/2 differential phase shift and DTD of either 10ps or 30ps. The responsivities of the balanced detector and the direct detector were assumed to be 0.6A/W and 0.9A/W respectively, and equivalent thermal noise spectral power densities were assumed to be 100pA/Hz

^{1/2}and 18pA/Hz

^{1/2}respectively. These receiver parameters match typical values of the commercial detectors used in the recent experimental demonstration [11]. After detection, the signals were electrically amplified, filtered by 15GHz 4

^{th}-order Bessel electrical filters (EFs), and down-sampled to 50Gsamples/s (5 samples per bit at 10Gbit/s) to simulate the sampling effect of the commonly implemented real-time oscilloscope. The down-sampled copy of

*V*

_{x}(

*t*) was re-biased, which can significantly enhance the robustness of the scheme to thermal noise. To allow for path length variations due to manufacturing tolerance, temperature variations and device aging, a sample shifting was included to provide delay tunability with a resolution equivalent to the sampling interval (20ps), and the amplitude was adjusted to correct for the scaling constant which equals 1/(2

*π*Δ

*t*) and any gain imbalance. As will be discussed later, additional signal processing stages included a phase compensator to significantly enhance the tolerance of differential phase misalignment of AMZI and a Gaussian-shaped high-pass EF to suppress the impairment from low-frequency amplification.

*V*

_{A}(

*t*) and

*V*

_{f}(

*t*) in Fig. 2 thus represent the recovered intensity and instantaneous frequency respectively. This information was exploited to reconstruct a replica of the optical signal, which was subsequently compensated using a dispersive transmission line. The compensated signal was then square-law detected and decoded using optimal threshold detection.

^{-4}by direct error counting. 128,961 bits were sufficient to produce a confidence interval of [3.5×10

^{-4}7×10

^{-4}] for this BER with 99% certainty [13

13. M. C. Jeruchim, “Techniques for estimating the bit error rate in the simulation of digital communication systems,” IEEE J. Sel. Areas Commun. **SAC-2**, 153–170 (1984). [CrossRef]

## 3. Suppression of low-frequency component amplification

*V*

_{p}(

*t*) to the estimated frequency

*V*

_{f}(

*t*):

*ψ*

_{p}(

*ω*) and

*ψ*

_{f}(

*ω*) are the spectra of the two electrical signals

*V*

_{p}(

*t*) and

*V*

_{f}(

*t*) respectively. It is clear from Eq.(4) that the low-frequency components of

*V*

_{f}(

*t*) dominate phase reconstruction, with a scaling factor of 2

*π*/|

*ω*|. Therefore, any noise or inaccuracy in the low-frequency components of the estimated

*V*

_{f}(

*t*) will accumulate and eventually limit the performance of the EDC. This suggests that the low-frequency components of

*V*

_{f}(

*t*) should be minimized.

*V*

_{x}(

*t*) were assumed to be 10ps and 0V respectively. Figure 3 shows

*ψ*

_{f}(

*ω*) [(a)-(c)] and eye diagrams of the signal after dispersion compensation [(d)-(f)] at a fiber length of 2160km under various conditions [(a) and (d): 25dB ER without a high-pass EF; (b) and (e): 12dB ER without a high-pass EF; (c) and (f): 12dB ER with a 0.85GHz high-pass EF]. For a larger ER [(a) and (d)], the received value of

*V*

_{A}(

*t*) for a sequence of consecutive logical data ‘0’s was so small that any optical noise led to large estimation inaccuracy in

*V*

_{f}(

*t*)=

*V*

_{y}(

*t*)/

*V*

_{A}(

*t*)

^{2}. This inaccuracy contained significant low-frequency content, which was further increased by the low-frequency component amplification. By using a smaller ER [(b) and (e)], the value of

*V*

_{A}(

*t*) for a sequence of consecutive logical data ‘0’s was increased, reducing the estimation inaccuracy of

*V*

_{f}. Furthermore, the positive and negative chirps of the dispersed optical signal were detected at all points in the signal, substantially reducing the low-frequency content of

*V*

_{f}. Consequently, the low-frequency component amplification in the phase reconstruction was also alleviated, leading to better compensation performance at a smaller ER, as shown in Fig. 3(e). The high-pass EF in the frequency estimation path further reduced the low-frequency components of

*V*

_{f}. As a result, the compensated OOK signal after 2160km transmission shown in Fig. 3(f) has a significantly clearer eye than that in Figs 3(d) and 3(e).

*V*

_{f}(

*t*). This distortion results in the rails of the eye diagrams in Fig. 3(f) being somewhat thicker than those in Fig. 3(e). Clearly, a trade-off exists between the impairment from low-frequency component amplification and the distortion. At an ER of 12dB, we optimized the bandwidth of the EF for 2160km transmission, which was around 0.85GHz as shown in Fig. 4(b). Note also that whilst a high-pass filter was placed in the path of

*V*

_{f}, no such filter was placed in the path of

*V*

_{A}, allowing the low-frequency components of the intensity signal

*V*

_{A}(

*t*) to pass through the compensator.

## 4. Fiber link performance

*V*

_{x}(

*t*) were assumed to be 10ps and 0V respectively, and photodiode thermal noise was neglected. From the figure, it is shown that including fiber nonlinearity in the transmission simulation resulted in an additional penalty of up to 1.8dB for system lengths less than 2160km. On the other hand, the maximum achievable OSNR degraded as the fiber length increased. Considering a 1550nm optical signal, 0.1nm (~12.5GHz) resolution and 4dB noise figure, for a total system length of 240km and 2160km (80km per span), the maximum achievable OSNRs were 30.2dB and 20.6dB respectively as shown by the squares in Fig. 5. In this figure, it is shown that at a fiber length of 2160km, the maximum achievable OSNR was more than 5 dB greater than the required value, confirming the significant improvement from the ER reduction and the suppression of low-frequency component amplification.

4. D. McGhan, M. O’Sullivan, M. Sotoodeh, A. Savchenko, C. Bontu, M. Belanger, and K. Roberts, “Electronic dispersion compensation,” in *Proc. Optical Fiber Communication Conference* (2006), paper OWK1. [CrossRef]

## 5. Enhanced robustness to thermal noise

*V*

_{x}(

*t*), to enhance the robustness of the scheme to thermal noise. Figure 7(b) shows the required OSNR versus the system length without (circles) and with (triangles) thermal noise with a 0.1

*M*bias added to

*V*

_{x}and 30ps AMZI DTD (

*M*represents the average detected signal amplitude ([

*E*{|

*V*

_{x}(

*t*)|

^{2}}]

^{1/2}) and

*E*{·} represents the ensemble average). The system performances using either a 0.1

*M*

*V*

_{x}(

*t*) bias or a 30ps AMZI DTD are also depicted in Fig. 7(b) by the squares and crosses respectively. From the figure, it is shown that without thermal noise, the proposed method had almost the same performance compared to the case of 0V

*V*

_{x}(

*t*) bias and 10ps AMZI DTD, as shown by the circles in Fig. 7. However, the method exhibited significantly greater tolerance to thermal noise, which we attribute to the improvement in the estimation of

*V*

_{f}(

*t*). Considering thermal noise alone,

*V*

_{y}(

*t*),

*V*

_{x}(

*t*), and

*V*

_{f}(

*t*) in Fig. 2 can be expressed as:

*n*

_{th_x}and

*n*

_{th_y}represent the thermal noise on

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*) respectively. The factor

*α*accounts both for the scaling constant, which is equal to 1/(2

*π*Δ

*t*), and for the potentially different gains of the

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*) signal paths. |

*E*(

*t*)|

^{2}is very small for logical data ‘0’s. From Eq. (5), it is clear that by employing an AMZI with a larger DTD, the output signal power from the balanced detector is increased since

*V*

_{y}(

*t*) is approximately proportional to Δ

*t*, which therefore improves the signal to thermal noise ratio of

*V*

_{y}(

*t*). Note that Δ

*t*cannot be increased infinitely because the principle for extraction of signal instantaneous frequency (Eq. (3)) is under the assumption of small Δ

*t*(Δ

*ω*(

*t*)Δ

*t*<<1). By adding a bias on

*V*

_{x}(

*t*), the resultant value of the denominator for sequences of consecutive logical data ‘0’s is increased, reducing the impact of thermal noise, although it causes slight distortions of

*V*

_{f}(

*t*) as well. In Fig. 8, the required OSNR versus normalized bias is plotted for 30ps AMZI DTD at a transmission length of 2160km with the total received optical powers of the balanced detector and direct detector equal to 0dBm. This figure shows that 0.1

*M*was a near-optimum value for combating thermal noise whilst maintaining the negligible

*V*

_{f}(

*t*) distortion. An alternative means to combat the effect of thermal noise is to simply increase the total power incident on the photodiodes, by increasing the preamplifier gain. Figure 9 shows the required OSNR versus the total received optical power at a system length of 2160km without thermal noise mitigation, and with various levels of mitigation: (circles: 0V bias and 10ps AMZI DTD; squares: 0.1

*M*bias and 10ps AMZI DTD; crosses: 0V bias and 30ps AMZI DTD; triangles: 0.1

*M*bias and 30ps AMZI DTD). The dashed line represents the reference case without thermal noise described above. From the figure, it is shown that without thermal noise mitigation, a received optical power larger than 9dBm was required to achieve tolerable penalty from the thermal noise. Such strict requirement could be significantly relaxed by either properly biasing

*V*

_{x}(

*t*) or using a AMZI with a larger DTD. By both biasing

*V*

_{x}(

*t*) and using an AMZI with a larger DTD (triangles), the required optical receiver power was reduced. As a result, less than 0.3dB thermal-noise induced OSNR penalty was achieved at 0dBm received optical power. This still enables sufficient OSNR margin (~4.8dB) for the transmission system, whilst also significantly relaxing the requirements for the gain of the preamplifier, the loss of the OBPF and the AMZI, and the power handling of the detectors.

## 6. Influence and compensation of variations in system parameters

*π*/2 and the paths resulting in

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*) were correctly matched. However, due to temperature variation and device aging, these parameters might drift over time, leading to additional OSNR penalty. Therefore, it is essential to investigate performance sensitivity to these parameters and to propose effective methods of compensation which can tolerate these kinds of drift. In addition, it is also desirable to determine the influence of the sampling rate to understand the trade-off between performance and electronic complexity. Note that in practice, full optical-field EDC may be implemented using analogue devices, in which case some parameters, e.g. influence of sampling rate, are not relevant. In the following simulations, we assume that both fiber nonlinearity and thermal noise are included.

*V*

_{x}(

*t*) has a 0.1

*M*bias and AMZI has a 30ps differential time delay.

### 6.1 Compensation of AMZI differential phase misalignment

*π*/2+Δ

*ζ*with Δ

*ζ*being a misalignment, we can derive from Fig. 2:

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*), which is included in Monte Carlo simulations, is not explicitly considered in Eq. (6). The first term of

*V*

_{f}(

*t*) in Eq. (6) estimates the instantaneous frequency (Δ

*ω*(

*t*)/(2

*π*)) with distortions due to added bias. The second term on the right-hand side of Eq. (6) which represents the impairment from the AMZI differential phase misalignment and is linearly proportional to the misalignment Δ

*ζ*, also introduces distortion to

*V*

_{f}(

*t*). The proposed AMZI differential phase compensator, included within the dotted box of Fig. 2, estimates the value of the second term and removes it from

*V*

_{f}(

*t*). To achieve this,

*V*

_{x}(

*t*) was processed to obtain |

*E*(

*t*)|

^{2}/(|

*E*(

*t*)|

^{2}+

*bias*) which was scaled by Δ

*ζ*/(2

*π*Δ

*t*) and subtracted from the estimated frequency prior to high pass filtering. An adaptive algorithm should be used to obtain an estimation of Δ

*ζ*in a practical system. Adaptive feedback might be implemented for example using forward error correction (FEC) error count or an eye monitor [15]. Due to slow phase variation, it is feasible to make the convergence speed of the algorithm faster than the variation speed of the phase drift. In this paper, to simplify the analysis, we adopted static compensation for each misalignment value.

*M*bias and (b) 0.3

*M*bias. The figure clearly shows that without the compensator, the performance of the full optical-field EDC was degraded by phase misalignment. The tolerance range at 1dB penalty was around ±120 in Fig. 10(a) for the optimal bias value of 0.1

*M*. This tolerance range was reduced to ±50 (with respect to the reference line with optimal bias and AMZI differential phase) by using a bias value of 0.3

*M*, which introduced larger distortion and consequently more OSNR penalty when the AMZI was detuned. By using the proposed method, the sensitivity to the differential phase misalignment was greatly suppressed for both bias values. The tolerance range was significantly increased up to ±240 and ±160 at 1dB penalty for bias of 0.1

*M*(Fig. 10(a)) and 0.3

*M*(Fig. 10(b)) respectively, which corresponded to more than doubled tolerance compared to those without the compensator.

### 6.2 Sampling synchronization

*V*

_{x}(

*t*) and

*V*

_{y}(

*t*), which may also change slowly with temperature and device aging. Such delay errors may be compensated using digitally implemented delays to a resolution equivalent to the sampling interval. This implies that the maximum OSNR penalty induced by poor synchronization may be estimated from the variation within this interval. Figure 11 shows the required OSNR versus delay error (

*D*

_{x}-(Δ

*t*/2+

*D*

_{y})) at a system length of 2160km, where

*D*

_{x}and

*D*

_{y}are the delays of the path

*V*

_{x}(

*t*) and the path which passed through the fast arm of the Δ

*t*-delay AMZI to result in

*V*

_{y}(

*t*) respectively. Δ

*t*was 30ps.

*V*

_{x}(

*t*) was assumed to be sampled with one in five samples occurring at maximum eye opening at 50Gsamples/s. The simulated required OSNR versus delay error is clearly asymmetric, but over the optimum 20ps window, the maximum OSNR penalty is an acceptable 0.7dB.

### 6.3 Effect of the sampling rate

## 7. Conclusions

## Acknowledgments

## References and links

1. | S. Schube and M. Mazzini, “Testing and interoperability of 10GBASE-LRM optical interfaces,” IEEE Commun. Mag. |

2. | F. Buchali, H. Bulow, W. Baumert, R. Ballentin, and T. Wehren, “Reduction of the chromatic dispersion penalty at 10Gbit/s by integrated electronic equalizers,” in |

3. | A. Farbert, S. Langenbach, N. Stojanovic, C. Dorschky, T. Kupfer, C. Schulien, J.-P. Elbers, H. Wernz, H. Griesser, and C. Glingener, “Performance of a 10.7 Gb/s receiver with digital equaliser using maximum likelihood sequence estimation,” |

4. | D. McGhan, M. O’Sullivan, M. Sotoodeh, A. Savchenko, C. Bontu, M. Belanger, and K. Roberts, “Electronic dispersion compensation,” in |

5. | G. Bosco and P. Poggiolini, “Long-distance effectiveness of MLSE IMDD receivers,” IEEE Photo. Technol. Lett. |

6. | M. G. Taylor, “Coherent detection for optical communications using digital signal processing,” in |

7. | A. D. Ellis and M. E. McCarthy, “Receiver-side electronic dispersion compensation using passive optical field detection for low cost 10Gbit/s 600 km-reach applications,” in |

8. | X. Liu, S. Chandrasekhar, and A. Leven, “Digital self-coherent detection”, Opt. Express |

9. | N. Kikuchi, K. Mandai, S. Sasaki, and K. Sekine, “Proposal and first experimental demonstration of digital incoherent optical field detector for chromatic dispersion compensation,” |

10. | A. Polley and S. E. Ralph, “Receiver-side adaptive opto-electronic chromatic dispersion compensation,” in |

11. | J. Zhao, M. E. McCarthy, P. Gunning, and A. D. Ellis, “Dispersion tolerance enhancement in electronic dispersion compensation using full optical-field reconstruction,” in |

12. | H. Haunstein and R. Urbansky, “Application of electronic equalization and error correction in lightwave systems,” |

13. | M. C. Jeruchim, “Techniques for estimating the bit error rate in the simulation of digital communication systems,” IEEE J. Sel. Areas Commun. |

14. | X. Liu and D. A. Fishman, “A fast and reliable algorithm for electronic pre-equalization of SPM and chromatic dispersion,” in |

15. | B. Franz, F. Buchali, D. Rosener, and H. Bulow, “Adaptation techniques for electronic equalizers for the mitigation of time-variant distortions in 43Gbit/s optical transmission systems,” in |

**OCIS Codes**

(060.2330) Fiber optics and optical communications : Fiber optics communications

(060.4080) Fiber optics and optical communications : Modulation

**ToC Category:**

Fiber Optics and Optical Communications

**History**

Original Manuscript: June 19, 2008

Revised Manuscript: July 25, 2008

Manuscript Accepted: July 25, 2008

Published: September 15, 2008

**Citation**

J. Zhao, M. E. McCarthy, and A. D. Ellis, "Electronic dispersion compensation using full optical-field reconstruction in 10Gbit/s OOK based systems," Opt. Express **16**, 15353-15365 (2008)

http://www.opticsinfobase.org/oe/abstract.cfm?URI=oe-16-20-15353

Sort: Year | Journal | Reset

### References

- S. Schube and M. Mazzini, "Testing and interoperability of 10GBASE-LRM optical interfaces," IEEE Commun. Mag. 45, s26-s31 (2007). [CrossRef]
- F. Buchali, H. Bulow, W. Baumert, R. Ballentin, and T. Wehren, "Reduction of the chromatic dispersion penalty at 10Gbit/s by integrated electronic equalizers," in Proc. Optical Fiber Communication Conference (2000), paper ThS1-1.
- A. Farbert, S. Langenbach, N. Stojanovic, C. Dorschky, T. Kupfer, C. Schulien, J.-P. Elbers, H. Wernz, H. Griesser, and C. Glingener, "Performance of a 10.7 Gb/s receiver with digital equaliser using maximum likelihood sequence estimation," European Conference on Optical Communication (2004), PDP Th4.1.5.
- D. McGhan, M. O'Sullivan, M. Sotoodeh, A. Savchenko, C. Bontu, M. Belanger, and K. Roberts, "Electronic dispersion compensation," in Proc. Optical Fiber Communication Conference (2006), paper OWK1. [CrossRef]
- G. Bosco and P. Poggiolini, "Long-distance effectiveness of MLSE IMDD receivers," IEEE Photo. Technol. Lett. 18, 1037-1039 (2006). [CrossRef]
- M. G. Taylor, "Coherent detection for optical communications using digital signal processing," in Proc. Optical Fiber Communication Conference (2007), paper OMP1.
- A. D. Ellis and M. E. McCarthy, "Receiver-side electronic dispersion compensation using passive optical field detection for low cost 10Gbit/s 600 km-reach applications," in Proc. Optical Fiber Communication Conference (2006), paper OTuE4.
- X. Liu, S. Chandrasekhar, and A. Leven, "Digital self-coherent detection", Opt. Express 16, 792-803 (2008). [CrossRef] [PubMed]
- N. Kikuchi, K. Mandai, S. Sasaki, and K. Sekine, "Proposal and first experimental demonstration of digital incoherent optical field detector for chromatic dispersion compensation," European Conference on Optical Communication (2006), PDP Th4.4.4.
- A. Polley and S. E. Ralph, "Receiver-side adaptive opto-electronic chromatic dispersion compensation," in Proc. Optical Fiber Communication Conference (2007), paper JThA51.
- J. Zhao, M. E. McCarthy, P. Gunning, and A. D. Ellis, "Dispersion tolerance enhancement in electronic dispersion compensation using full optical-field reconstruction," in Proc. Optical Fiber Communication Conference (2008), paper OWL3.
- H. Haunstein and R. Urbansky, "Application of electronic equalization and error correction in lightwave systems," European Conference on Optical Communication (2004), paper Th1.5.1.
- M. C. Jeruchim, "Techniques for estimating the bit error rate in the simulation of digital communication systems," IEEE J. Sel. Areas Commun. SAC-2, 153-170 (1984). [CrossRef]
- X. Liu and D. A. Fishman, "A fast and reliable algorithm for electronic pre-equalization of SPM and chromatic dispersion," in Proc. Optical Fiber Communication Conference (2006), paper OThD4.
- B. Franz, F. Buchali, D. Rosener, and H. Bulow, "Adaptation techniques for electronic equalizers for the mitigation of time-variant distortions in 43Gbit/s optical transmission systems," in Proc. Optical Fiber Communication Conference (2007), paper OMG1.

## Cited By |
Alert me when this paper is cited |

OSA is able to provide readers links to articles that cite this paper by participating in CrossRef's Cited-By Linking service. CrossRef includes content from more than 3000 publishers and societies. In addition to listing OSA journal articles that cite this paper, citing articles from other participating publishers will also be listed.

### Figures

Fig. 1. |
Fig. 2. |
Fig. 3. |

Fig. 4. |
Fig. 5. |
Fig. 6. |

Fig. 7. |
Fig. 8. |
Fig. 9. |

Fig. 10. |
Fig. 11. |
Fig. 12. |

« Previous Article | Next Article »

OSA is a member of CrossRef.