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Optics Express

Optics Express

  • Editor: C. Martijn de Sterke
  • Vol. 20, Iss. 9 — Apr. 23, 2012
  • pp: 10339–10352
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Towards full band colorless reception with coherent balanced receivers

Bo Zhang, Christian Malouin, and Theodore J. Schmidt  »View Author Affiliations


Optics Express, Vol. 20, Issue 9, pp. 10339-10352 (2012)
http://dx.doi.org/10.1364/OE.20.010339


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Abstract

In addition to linear compensation of fiber channel impairments, coherent receivers also provide colorless selection of any desired data channel within multitude of incident wavelengths, without the need of a channel selecting filter. In this paper, we investigate the design requirements for colorless reception using a coherent balanced receiver, considering both the optical front end (OFE) and the transimpedance amplifier (TIA). We develop analytical models to predict the system performance as a function of receiver design parameters and show good agreement against numerical simulations. At low input signal power, an optimum local oscillator (LO) power is shown to exist where the thermal noise is balanced with the residual LO-RIN beat noise. At high input signal power, we show the dominant noise effect is the residual self-beat noise from the out of band (OOB) channels, which scales not only with the number of OOB channels and the common mode rejection ratio (CMRR) of the OFE, but also depends on the link residual chromatic dispersion (CD) and the orientation of the polarization tributaries relative to the receiver. This residual self-beat noise from OOB channels sets the lower bound for the LO power. We also investigate the limitations imposed by overload in the TIA, showing analytically that the DC current scales only with the number of OOB channels, while the differential AC current scales only with the link residual CD, which induces high peak-to-average power ratio (PAPR). Both DC and AC currents at the input to the TIA set the upper bounds for the LO power. Considering both the OFE noise limit and the TIA overload limit, we show that the receiver operating range is notably narrowed for dispersion unmanaged links, as compared to dispersion managed links.

© 2012 OSA

1. Introduction

As data and video traffic continue to grow with no end in sight, service providers feel great pressure to deploy reconfigurable optical network architectures to enable flexible and efficient service provisioning in order to address unpredictable traffic demand. At the center of this versatility is reconfigurable optical add/drop multiplexers (ROADMs) [1

1. M. D. Feuer, D. C. Kilper, and S. L. Woodward, “ROADMs and their system applications,” in Optical Fiber Telecommunications, V, I. Kaminow, T. Li and A. E. Willner, eds., (Elsevier, 2008) Vol. B. Chap. 8, 293–344.

], which is capable of routing each data channel independently at any node. In today’s network, colored optical demultiplexers are used, where optical filters at fixed wavelengths provide the optical frequency discrimination required for direct detection receivers. A growing trend is to employ “colorless” ROADM architectures, where the ROADM broadcasts a given number of optical channels to the drop path and a coherent receiver selects the optical channel of interest from the rest. This enables any wavelength to be flexibly routed to any receiver without manual intervention, simplifying provisioning of new services and opening the door for backup protection against transceiver failure as well as bandwidth-on-demand services. All these crucial applications require an optical receiver that can, without the need of an optical filter, detect any wavelength channel out of a number of incident DWDM channels.

The fact that an optical demultiplexer is not necessary when coherent detection is employed has been known for more than two decades [2

2. E.-J. Bachus, R.-P. Braun, C. Caspar, E. Grossmann, H. Foisel, K. Heimes, H. Lamping, B. Strebel, and F.-J. Westphal, “Ten-channel coherent optical fibre transmission,” Electron. Lett. 22(19), 1002–1003 (1986). [CrossRef]

]. With coherent receivers, channel selection can be achieved simply by tuning the local oscillator (LO) close to the desired channel and the filtering is achieved in the baseband to suppress any mixed signal-signal beat notes. Balanced receivers have also long been recognized to enable homodyne detection and to allow tight channel spacing, with the distinct advantage of further suppressing the signal self-beat common mode noise [3

3. L. Kazovsky, “Multichannel coherent optical communications systems,” J. Lightwave Technol. 5(8), 1095–1102 (1987). [CrossRef]

]. However, the lack of high-speed mixed signal circuits, the difficulty of making an ideal balanced photo-detector, and a lack of commercially viable narrow linewidth LO lasers two decades ago, limited the practical understanding of the intricate receiver design tradeoffs, and, to certain extent, resulted in a shift of research focus to optically amplified systems with direct detection. Recently, largely due to the success of commercializing digital signal processing (DSP) technologies, coherent detection, specifically with 40-Gb/s and 100-Gb/s polarization-multiplexed quadrature phase shift keying (PM-QPSK) optical modulation formats, have proven to be real and ready for commercial deployment [4

4. H. Sun, K. T. Wu, and K. Roberts, “Real-time measurements of a 40 Gb/s coherent system,” Opt. Express 16(2), 873–879 (2008). [CrossRef] [PubMed]

, 5

5. M. Birk, P. Gerard, R. Curto, L. E. Nelson, X. Zhou, P. Magill, T. J. Schmidt, C. Malouin, B. Zhang, E. Ibragimov, S. Khatana, M. Glavanovic, R. Lofland, R. Marcoccia, R. Saunders, G. Nicholl, M. Nowell, and F. Forghieri, “Real-time single-carrier coherent 100 Gb/s PM-QPSK field trial,” J. Lightwave Technol. 29(4), 417–425 (2011). [CrossRef]

]. For colorless reception, recent literature reports the use of single-ended receiver designs, which either suffer from limited channel count and very poor dynamic range [6

6. L. E. Nelson, S. L. Woodward, S. Foo, M. Moyer, D. J. S. Beckett, M. O’Sullivan, and P. D. Magill, “Detection of a single 40 Gb/s polarization-multiplexed QPSK channel with a real-time intradyne receiver in the presence of multiple coincident WDM channels,” J. Lightwave Technol. 28(20), 2933–2943 (2010). [CrossRef]

], or require the involvement of a more complicated receiver front-end [7

7. C. Xie, P. J. Winzer, G. Raybon, A. H. Gnauck, B. Zhu, T. Geisler, and B. Edvold, “Colorless coherent receiver using 3x3 coupler hybrids and single-ended detection,” in Proceedings of ECOC, postdeadline paper, Th.13.b.2, (2011).

]. We note that with the maturity of integrated photonic technologies [8

8. C. R. Doerr, L. Zhang, P. J. Winzer, N. Weimann, V. Houtsma, T. Hu, N. J. Sauer, L. L. Buhl, D. T. Neilson, S. Chandrasekhar, and Y. K. Chen, “Monolithic InP dual-polarization and dual-quadrature coherent receiver,” IEEE Photon. Technol. Lett. 23(11), 694–696 (2011). [CrossRef]

] and volume production, the cost of balanced detection receivers is anticipated to closely approach that of single-ended versions.

2. Noise sources in a colorless coherent receiver

We first discuss the noise terms unique to the colorless application. Here, OOB is defined as wavelength bands other than the data channel under detection. As shown in Fig. 1, when multiple channels are presented to the coherent intradyne receiver, the interference term at baseband is not due to the LO beating with the OOB channels, nor the OOB channels beating with each other, because those mixing products are out of the baseband detection bandwidth. The interference term is due to each of the OOB channel mixing with itself. We call this interference term “OOB self-beat noise”. Signal-ASE beat noise and ASE-ASE beat noise from the OOB channels will also show up in the receiver, but the strength is much smaller when we consider similar occupied bandwidth for both ASE and the OOB channels. As discussed in [6

6. L. E. Nelson, S. L. Woodward, S. Foo, M. Moyer, D. J. S. Beckett, M. O’Sullivan, and P. D. Magill, “Detection of a single 40 Gb/s polarization-multiplexed QPSK channel with a real-time intradyne receiver in the presence of multiple coincident WDM channels,” J. Lightwave Technol. 28(20), 2933–2943 (2010). [CrossRef]

], if we use the per channel average power squared to represent this “OOB self-beat noise” variance, a scaling factor, which takes into account the detailed signal-signal beat spectrum, must be present. This scaling factor is a function of the residual CD and the orientation of the polarization tributaries of the OOB channel at the input to the receiver. This could be understood from the fact that both CD and polarization orientation will increase the PAPR of each OOB data signal [6

6. L. E. Nelson, S. L. Woodward, S. Foo, M. Moyer, D. J. S. Beckett, M. O’Sullivan, and P. D. Magill, “Detection of a single 40 Gb/s polarization-multiplexed QPSK channel with a real-time intradyne receiver in the presence of multiple coincident WDM channels,” J. Lightwave Technol. 28(20), 2933–2943 (2010). [CrossRef]

], and thus enhance the beat-noise variance. On the other hand, these interference terms from the OOB channels are common to both photodiodes in a balanced detector and are thus greatly suppressed inside the differential TIA. The suppression ratio is quantified by the CMRR of the OFE [9

9. Y. Painchaud, M. Poulin, M. Morin, and M. Têtu, “Performance of balanced detection in a coherent receiver,” Opt. Express 17(5), 3659–3672 (2009). [CrossRef] [PubMed]

, 10

10. B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

].

We then briefly discuss the intrinsic noise terms associated with the coherent balanced receiver [3

3. L. Kazovsky, “Multichannel coherent optical communications systems,” J. Lightwave Technol. 5(8), 1095–1102 (1987). [CrossRef]

, 10

10. B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

]. These noise terms will dictate the low end of the input signal dynamic range.

Shot noise is a manifestation of the fact that an electric current consists of a stream of electrons that are generated at random times. Four sets of balanced receivers independently add uncorrelated shot noise of their own onto the overall incident signal. In colorless receivers, it will scale will the number of OOB channels, as well as the LO power. For a nominal LO power to signal power ratio, the amount of excess shot noise is relatively small compared to other noise terms. The dark current is usually small and can thus be neglected.

Thermal noise is due to random thermal motion of electrons in a resistor, manifesting as a fluctuating current even in the absence of an applied voltage. Usually, noise generated by the transimpedance amplifiers (TIAs) is the dominating thermal noise term and the amount added depends on the front-end design. A simple approach to account for the thermal noise is to use the equivalent TIA input-referred noise current density [10

10. B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

].

Another important noise source comes from the relative intensity noise (RIN) of the LO. From each photodiode’s perspective, the beating between the LO and its intensity noise acts as an interference term onto the useful signal-LO beat term. Balanced detection will suppress this term by a finite rejection ratio. The residual will leak through and act as a non-negligible interference noise term, especially when the LO power is high.

3. Effective CMRR in a coherent balanced receiver

When the received signal and the LO are equally split and synchronized from the last 2x2 coupler inside the 90 degree hybrid to each set of balanced PD, perfect cancellation of the common mode can be achieved. In reality, power imbalance and skew mismatch leads to a finite CMRR. Power imbalance can result from deviations in a 50/50 splitting ratio from the last coupler as well as unequal responsivities of the PDs. Skew mismatches are due to the P and N path length differences from the output of the last 2x2 coupler to the two inputs of the differential TIA. If we define the CMRR as the power ratio of the residual common mode with respect to the combined common mode [11

11. OIF IA # OIF-DPC-RX-01.0, “Implementation agreement for integrated dual polarization intradyne coherent receivers,” April 16, 2010.

], and translate that into the frequency domain, one can write the CMRR of a balanced mixer as follows [10

10. B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

],
CMRR(f)=|α2ej2πfτα2+1|2
(1)
where αis the power imbalance in linear scale including both the unequal splitting ratio as well as unequal PD responsivity, and τis the P/N skew of the four tributaries. One can see from the above equation that at low frequencies, CMRR is dictated by the power imbalance, whereas the P/N skew determines the frequency dependent behavior of the CMRR [9

9. Y. Painchaud, M. Poulin, M. Morin, and M. Têtu, “Performance of balanced detection in a coherent receiver,” Opt. Express 17(5), 3659–3672 (2009). [CrossRef] [PubMed]

].

Due to the frequency dependent nature of the CMRR, there is a need to introduce the concept of “effective CMRR” as a single averaged factor which can be applied to residual beat noise terms. To do so, we show in Fig. 2(a)
Fig. 2 (a) CMRR as a function of frequency for various P/N skew levels and P/N power imbalance values. The normalized one-sided 32-Gbaud PM-QPSK optical spectrum is also overlaid. (b) Effective CMRR value based on different combinations of P/N skew values and P/N power imbalance values. The effective CMRR is extracted from (a) at around 8 to 10 GHz frequency content.
the one-sided optical spectrum of a 32-Gbaud PM-QPSK signal. The spectrum notches at the baud frequency of ~32GHz. The Nyquist frequency is defined as half the symbol rate, which is ~16GHz for our case. At 10GHz offset, the signal is attenuated by ~3dB with respect to the carrier wavelength. Based on Eq. (1), we plot the CMRR as a function of frequency for various combinations of P/N skew and P/N power imbalance. Based on the 16-20GHz nominal receiver analog bandwidth as well as the frequency dependent shape (depends on P/N power imbalance and P/N skew) of the CMRR curve, we choose CMRR at 8-10GHz (around half the Nyquist frequency) as the “effective CMRR” value. Although the effective CMRR does not give an exact account of the total common mode noise, it provides the effect of the residual noise and allows one to make quantitative predictions on the receiver performance. A mapping of a combination of skew and power imbalance to effective CMRR value is thus shown in Fig. 2(b). Here the P/N power imbalance is shown as 10*log10(α)in dB. For instance, 0.5dB corresponds to a 53% to 47% power splitting imbalance, which could be achieved in a typical 2x2 coupler.

4. Theory for performance of a coherent colorless receiver

After laying out the various noise sources in a colorless receiver and the definition of the “effective CMRR”, we analytically derive the performance bounds by examining the dominant noise sources under different operating scenarios. The currents at port P and port N of any of the four balanced photo-detectors can be represented as follows,
Ip(t)=|(ESIG,s(t)+EASE,s)+i=1N(ESIG,i(t)+EASE,i)+(ELO+ELOIN(t))|2+iTIA+ishot
(2)
In(t)=α|(ESIG,s(t+τ)+EASE,s)+i=1N(ESIG,i(t+τ)+EASE,i)+(ELO+ELOIN(t+τ))|2+iTIA+ishot
(3)
whereESIG,s, ESIG,i,EASE,s,EASE,i,ELO, and ELOINare the electric fields of the signal under detection, the ith signal out of a number N of OOB signals, ASE of the signal, ASE of the ith signal, LO and LO intensity noise, measured at the input to the PD, respectively. Note that the noise-free ELOcontains constant amplitude, whereas the LO intensity noise ELOINbears the time-varying amplitude noise. τrepresents the differential delay between the P and N ports, whereas α represents the P/N power imbalance level. Both the signal and the LO intensity noise on the P port is only a time (skew) shifted version of the ones on the N port and thus are correlated with each other. iTIA is the thermal noise current from the TIA, and ishotis the shot noise current from the PD. Both of these two noise terms on the P and N ports are uncorrelated and thus will be doubled after balanced detection.

The SNR degradation (from the input to the output of the OFE) can be quantified by examining the strength of the additive noise terms except for the first term, signal to interference ratio (SIR) is introduced below to capture this SNR degradation. Note that here the “interference” in SIR is generally referred to as both the interference from OOB channels as well as the noise sources from the receiver.

SIR=PLO*PSIG,sσthermal2+σshot2+CMRR*[PLO*PLOIN+i=1N(|PASE,i|2+PASE,i*PSIG,i+β|PSIG,i|2]
(5)

Equation (5) can be simplified to provide insight if we consider two extremes of the input dynamic range. For low input power, i.e., −20dBm per channel, SIR can be represented by

SIR=PLO*PSIG,sσthermal2+σshot2+CMRR*PLO*PLOIN
(6)

Based on the analysis in an earlier publication of ours [10

10. B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

], and mainly due to the quadratic dependence of LO power from the LO-RIN beat noise term, an optimum LO power exists which maximizes the SIR and thus in turn minimizes the performance degradation.

On the high-end of the input power, i.e., 0dBm/channel, SIR can be simplified as,

SIR=PLOCMRR*β*N*PSIG,i
(7)

An even simpler equation can be provided if we convert Eq. (7) from linear scale to logrithmic scale,

SIR=LSRCMRR10*log(β*N)
(8)

Equation (8) reveals that as long as one has a large LO to signal power (per channel) ratio (LSR) or a good balanced detector (high CMRR value), there is potential to support a large number of OOB channel, without sacrificing too much the SIR value, and thus maintaining a reasonable system performance. Based on [6

6. L. E. Nelson, S. L. Woodward, S. Foo, M. Moyer, D. J. S. Beckett, M. O’Sullivan, and P. D. Magill, “Detection of a single 40 Gb/s polarization-multiplexed QPSK channel with a real-time intradyne receiver in the presence of multiple coincident WDM channels,” J. Lightwave Technol. 28(20), 2933–2943 (2010). [CrossRef]

], the scaling factor,β, depending on the amount of CD and the orientation of the polarization with respect to the fixed polarization axis of the PBS, ranges from 0.05 to 0.55. This range corresponds to a variation of 10-11dB in power, which matches the PAPR variation ranges caused by the combination of CD and polarization orientation. In our analytical simulations, we derive the dBQ penalty out of the SNR penalty shown above assuming Gaussian distribution of overall noise sources.

5. OFE Numerical simulations and analytical predictions

In order to validate the above analytical model, numerical simulations are carried out in Matlab. To generate the transmitter, 126.5 Gb/s (31.625 GBaud) non-return-to-zero polarization multiplexed quadrature phase shift keying (NRZ-PM-QPSK) modulated signal is prepared with up to 219-1 pseudo random bit sequence (PRBS) length on each of the four tributaries with different seeds. OOB channels ranging from 0 to 80 wavelengths (total number of channels from 1 to 81 channels) are generated with the same bit rate and modulation format at 50GHz spacing. To mimic the statistically uncorrelated multi-channel transmitter in a practical system, diverse PRBS seeds and different timing delays are used to randomize each channel. Orientation of the polarization tributaries for each channel are also varied based on the Jones angle and ellipticity with respect to the receiver PBS. DWDM signals are transmitted through various net CD values ranging from 0ps/nm to 50ns/nm, representing dispersion managed and unmanaged (DCM-free) links, respectively.

On the receiver side, a laser with 100 kHz linewidth and a RIN value of −145 dB/Hz is used for the local oscillator (LO), with an output power swept from 0 dBm to 20 dBm. The two 90 degree hybrids are modeled to have perfect I/Q and X/Y power balance, whereas the P/N skew and P/N power imbalance represents the source of imperfection, which contribute to an effective CMRR ranging from −13dB to −25dB. Excess loss from the hybrid is assumed to be 2 dB and the PD responsivity is set at 0.6 A/W. Thermal noise is modeled to have a differential TIA input noise current density of 18.2 pA/sqrt(Hz). The overall receiver analog bandwidth is modeled as having a 3dB bandwidth of 16GHz with a 5th order Butterworth shape. The two samples per symbol analog-to-digital converter (ADC) has an effective number of bits (ENOB) of 6, which is close to ideal for PM-QPSK signals. A 2x2 butterfly adaptive equalizer is realized in the frequency domain using least mean square (LMS) algorithm. Viterbi-viterbi based carrier phase estimation is performed to compensate for the phase noise. Error counting is performed to calculate the bit-error-rate (BER) and then converted to Q2 factor to quantify the system performance. Figure 3
Fig. 3 Key parameters with brief description used in the numerical simulation.
summarizes the key parameters used in our numerical simulations.

5.1 At low input signal power

Figure 4(b) shows that for a nominal LO power of 13dBm and an effective CMRR of −19dB (PN skew of 4ps), the system performance is largely independent of the number of OOB channels when the input power per channel is low (e.g. −20dBm). This can be explained by the fact that the residual self-beat noise from the OOB channels is negligible compared to the LO power, and thus contributes little additional interference noise to the ASE noise floor. The numerical simulation shows a slightly higher penalty (0.2-0.3dB) than the analytical predictions. This dependence can be attributed to the beating between the sidebands of the adjacent channel pairs which build up slightly when the OOB channel count increases.

5.2 At high input signal power

When the input power per channel is strong, i.e. 0dBm/channel, the receiver performance is expected to degrade with increasing number of OOB channels. However, as predicted by Eq. (8), increasing the LO power or improving the CMRR value can both significantly minimize the penalty. We show in Fig. 5(a)
Fig. 5 (a) Numerical and analytical simulations of Q2 penalty as a function of number of OOB channels for various LO powers. (b) Numerical and analytical simulations of Q2 penalty as a function of OOB channel number for various P/N skew values. The numerical simulation and analytical equations are seen to be in good agreement.
the Q2 penalty versus the number of OOB channels for different LO power, for a fixed signal power per channel of 0dBm and an effective CMRR of −19dB for a P/N skew of 4ps. The overall CD is 50ns/nm to represent a dispersion uncompensated (DCM-free) link, corresponding to a beta factor of 0.55. For a given number of OOB channels, higher LO power helps suppress the “residual self-beat noise” and thus improve the system performance. When the OOB channel count is 40, increasing the LO power from 7dBm to 13dBm reduces the Q2 penalty by approximately 1.5dB at an OSNR of 16dB/0.1nm.

Figure 5(b) shows the impact of P/N skews on the performance of colorless reception. For 40 OOB channels, reducing the skew from 8ps to 2ps (equating to a CMRR improvement of 12dB) improves system performance by close to 3.5dBQ. As evident from Eq. (8), both CMRR and LO power contribute on the same scale to the SIR. Figure 5(a) and 5(b) show very good agreement between our numerical simulations and analytical models.

5.3 Enhancing the input dynamic range

To further illustrate the impact of LO power and CMRR on the performance of colorless reception, Fig. 6(b) shows the simulation results when the OOB channels are set to 40. The inner three curves corresponds to an effective CMRR of −19dB (P/N skew of 4ps), while the LO power is changed from 13dBm to 19dBm. We see that increasing the LO power enlarges the high-end input power limit (thanks to a higher LSR value), but sacrifices the low-end input power limit due to a higher residual LO-RIN beat noise. On the other hand, the purple curve shows the benefit of enhancing the CMRR on both ends of the input dynamic range. For a fixed 13dBm LO power, reducing the P/N skew from 4 to 2ps helps increase the dynamic range by close to 10dB (4dB improvement on the low-end and 6dB improvement on the high-end). This can be explained by a higher CMRR helping reduce the residual LO-RIN beat noise as well as the residual self-beat noise from the OOB channels. The analytical simulations are seen to be in good agreement with numerical simulations.

5.4 Impact of polarization orientation with and without CD

To analytically predict the performance variation at higher OOB channel count, Fig. 7(b) shows the Q2 penalty versus the input signal power for an 80 OOB channel system. LO power is set to 13dBm and the effective CMRR is −19dB (P/N skew is 4ps). We see similar variation in terms of polarization orientation when there is no CD in the link. ~0.87dBQ variation is predicted when the signal power is 0dBm/channel. With CD in the link, the performance is not impacted by the orientation angle, as the PAPR and thus the self-beat noise variance saturates to a fixed level.

6. TIA overload and performance bounds

7. Discussion

8. Conclusion

Acknowledgment

We would like to acknowledge Keith Nellis, Hari Shankar, and Tom Broekaert from InPhi for useful discussions on the TIA overload topic. We would also like to thank the anonymous reviewers for their comments and suggestions.

References and links

1.

M. D. Feuer, D. C. Kilper, and S. L. Woodward, “ROADMs and their system applications,” in Optical Fiber Telecommunications, V, I. Kaminow, T. Li and A. E. Willner, eds., (Elsevier, 2008) Vol. B. Chap. 8, 293–344.

2.

E.-J. Bachus, R.-P. Braun, C. Caspar, E. Grossmann, H. Foisel, K. Heimes, H. Lamping, B. Strebel, and F.-J. Westphal, “Ten-channel coherent optical fibre transmission,” Electron. Lett. 22(19), 1002–1003 (1986). [CrossRef]

3.

L. Kazovsky, “Multichannel coherent optical communications systems,” J. Lightwave Technol. 5(8), 1095–1102 (1987). [CrossRef]

4.

H. Sun, K. T. Wu, and K. Roberts, “Real-time measurements of a 40 Gb/s coherent system,” Opt. Express 16(2), 873–879 (2008). [CrossRef] [PubMed]

5.

M. Birk, P. Gerard, R. Curto, L. E. Nelson, X. Zhou, P. Magill, T. J. Schmidt, C. Malouin, B. Zhang, E. Ibragimov, S. Khatana, M. Glavanovic, R. Lofland, R. Marcoccia, R. Saunders, G. Nicholl, M. Nowell, and F. Forghieri, “Real-time single-carrier coherent 100 Gb/s PM-QPSK field trial,” J. Lightwave Technol. 29(4), 417–425 (2011). [CrossRef]

6.

L. E. Nelson, S. L. Woodward, S. Foo, M. Moyer, D. J. S. Beckett, M. O’Sullivan, and P. D. Magill, “Detection of a single 40 Gb/s polarization-multiplexed QPSK channel with a real-time intradyne receiver in the presence of multiple coincident WDM channels,” J. Lightwave Technol. 28(20), 2933–2943 (2010). [CrossRef]

7.

C. Xie, P. J. Winzer, G. Raybon, A. H. Gnauck, B. Zhu, T. Geisler, and B. Edvold, “Colorless coherent receiver using 3x3 coupler hybrids and single-ended detection,” in Proceedings of ECOC, postdeadline paper, Th.13.b.2, (2011).

8.

C. R. Doerr, L. Zhang, P. J. Winzer, N. Weimann, V. Houtsma, T. Hu, N. J. Sauer, L. L. Buhl, D. T. Neilson, S. Chandrasekhar, and Y. K. Chen, “Monolithic InP dual-polarization and dual-quadrature coherent receiver,” IEEE Photon. Technol. Lett. 23(11), 694–696 (2011). [CrossRef]

9.

Y. Painchaud, M. Poulin, M. Morin, and M. Têtu, “Performance of balanced detection in a coherent receiver,” Opt. Express 17(5), 3659–3672 (2009). [CrossRef] [PubMed]

10.

B. Zhang, C. Malouin, and T. J. Schmidt, “Design of coherent receiver optical front end for unamplified applications,” Opt. Express 20(3), 3225–3234 (2012). [CrossRef] [PubMed]

11.

OIF IA # OIF-DPC-RX-01.0, “Implementation agreement for integrated dual polarization intradyne coherent receivers,” April 16, 2010.

12.

B. Razavi, Design of Integrated Circuits for Optical Communication Systems (McGraw-Hill, 2003).

OCIS Codes
(060.0060) Fiber optics and optical communications : Fiber optics and optical communications
(060.2380) Fiber optics and optical communications : Fiber optics sources and detectors

ToC Category:
Fiber Optics and Optical Communications

History
Original Manuscript: March 14, 2012
Revised Manuscript: April 8, 2012
Manuscript Accepted: April 12, 2012
Published: April 19, 2012

Citation
Bo Zhang, Christian Malouin, and Theodore J. Schmidt, "Towards full band colorless reception with coherent balanced receivers," Opt. Express 20, 10339-10352 (2012)
http://www.opticsinfobase.org/oe/abstract.cfm?URI=oe-20-9-10339


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References

  1. M. D. Feuer, D. C. Kilper, and S. L. Woodward, “ROADMs and their system applications,” in Optical Fiber Telecommunications, V, I. Kaminow, T. Li and A. E. Willner, eds., (Elsevier, 2008) Vol. B. Chap. 8, 293–344.
  2. E.-J. Bachus, R.-P. Braun, C. Caspar, E. Grossmann, H. Foisel, K. Heimes, H. Lamping, B. Strebel, and F.-J. Westphal, “Ten-channel coherent optical fibre transmission,” Electron. Lett.22(19), 1002–1003 (1986). [CrossRef]
  3. L. Kazovsky, “Multichannel coherent optical communications systems,” J. Lightwave Technol.5(8), 1095–1102 (1987). [CrossRef]
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